There are several power converter topologies that have been developed over the years, which are intended to improve the power density and switching efficiency of power converters. An emerging focus of new converter topologies is to provide a means to reduce or eliminate converter switching losses, while increasing the switching frequencies. Lower loss and higher switching frequency means more efficient converters, which can reduce the size and weight of converter components. Additionally, with the introduction of high speed composite semiconductor switches, such as metal oxide semiconductor field effect transistor (MOSFET) switches operated by pulse width modulation (PWM), recent forward and flyback topologies are now capable of operation at greatly increased switching frequencies, such as, for example, up to 1.0 MHz.
However, an increase in switching frequency can cause a corresponding increase in switching and component stress related losses, as well as increased electromagnetic interference (EMI), noise, and switching commutation problems, due to the rapid switching of the semiconductor switches at high voltage and/or high current levels. Moreover, modern electronic components are expected to perform multiple functions, in a small space, efficiently, and with minimal undesirable side effects. For instance, a conventional voltage converter that provides for relatively high power density and high switching frequencies, should also include uncluttered circuit topologies, provide for isolation of the output or “load” voltage from the input or “source” voltage, and also provide for variable step-up or step-down voltage transformation. In an effort to reduce or eliminate the switching losses and reduce EMI noise the use of “resonant” or “soft” switching techniques has been increasingly employed in the art. The application of resonant switching techniques to conventional power converter topologies offers many advantages for high density, and high frequency, to reduce or eliminate switching stress and reduce EMI. However, the complexity required to provide control to the power switches and the components associated with complex control, create a limited use in commercial applications.
In an effort to reduce or eliminate the switching losses and reduce EMI noise caused by high switching frequencies, “resonant” or “soft” switching techniques are increasingly being employed. Resonant switching techniques generally include an inductor-capacitor (LC) subcircuit in series with a semiconductor switch which, when enabled, creates a resonating subcircuit within the converter. Further, timing the control cycles of the resonant switch to correspond with particular voltage and current conditions across respective converter components during the switching cycle allows for switching under zero voltage and/or zero current conditions. Zero voltage switching (ZVS) and/or zero current switching (ZCS) inherently reduces or eliminates many frequency related switching losses. Several power converter topologies have been developed utilizing resonant switching techniques, such as, for example, U.S. Pat. No. 5,694,304 entitled “High Efficiency Resonant Switching Converters,” to Telefus, et al., (Telefus), which is hereby incorporated by reference; U.S. Pat. No. 5,057,986 entitled “Zero Voltage Resonant Transition Switching Power Converter,” to Henze, et al., (Henze), which is hereby incorporated by reference; U.S. Pat. No. 5,126,931 entitled “Fixed Frequency Single Ended Forward Converter Switching at Zero Voltage,” to Jitaru (Jitaru), which is hereby incorporated by reference; and U.S. Pat. No. 5,177,675 entitled “Zero Voltage, Zero Current, Resonant Converter,” to Archer, (Archer), which is hereby incorporated by reference.
In particular, Henze describes single ended DC-DC flyback topologies for operation at very high switching frequencies, such as 1.0 MHz or greater. In Henze, a plurality of pulse width modulated (PWM) switches are utilized to effect zero voltage resonant transition switching. Jitaru describes variations of known forward and/or flyback converter topologies employing zero voltage and/or zero current resonant techniques. Jitaru specifically describes a forward converter topology utilizing resonant switching techniques to operate at constant frequency. Archer describes zero voltage, and zero current, switching techniques in resonant flyback topologies utilizing a resonant transformer assembly inserted in parallel with either the primary or secondary winding of the main transformer.
The application of such resonant switching techniques to conventional power converter topologies offers many advantages for high density, high frequency converters, such as quasi sinusoidal current waveforms, reduced or eliminated switching stresses on the electrical components of the converter, reduced frequency dependent losses, and/or reduced EMI. However, energy losses incurred during control of zero voltage switching and/or zero current switching, and losses incurred during driving, and controlling the resonance means, are still problematic. For instance, some researchers have implemented an active clamp in conjunction with a resonant converter circuit to realize the benefits of high frequency switching, while reducing its many side effects. See, for example, the United States Patent to Telefus, incorporated by reference above.
An improved switching type power converter known as a quasi resonant tank circuit is described in US. Patent Publication No. 2007-0263415 to Jansen et. al. (Jansen), hereby incorporated in its entirety. FIG. 1 shows such a power converter 100 having a quasi resonant tank circuit 101 in simplified form. The circuit of FIG. 1 illustrates a conceptual representation of the invention in a Quasi Resonant Flyback converter. The power converter 100 comprises an output transformer 103 with primary and secondary windings, a primary switch 105, an auxiliary switch 104, a first resonance capacitor 106, a second resonance capacitor 102 and a comparator 109 with driving means for the auxiliary switch 104. The converter further includes secondary rectifier means comprising a diode 107 and a reservoir capacitor 108. In this exemplary embodiment, the primary switch 105 is controlled by a primary control module 111. The circuit in FIG. 1 includes a DC power source 112 to provide power to the primary side of the power converter. The comparator 109 and driver means for auxiliary switch 104 are configured such that when the voltage across the primary winding of the transformer 103 is higher than zero, the auxiliary switch will be enabled, or conducting. The comparator 109 and driver means for auxiliary switch 104 are further configured such that when the voltage across the primary winding of the transformer 103 is equal or lower than zero, the auxiliary switch will be disabled. Consequently a first resonance frequency exists for voltages of less than or equal to zero across the primary winding of transformer 103 as a result of the energy exchange between the primary inductance of transformer 103 and the first resonance capacitor 106.
It is well known that MOSFET switches include inherent capacitances that must be accounted for. To that end, FIG. 2 shows the power converter 100 of FIG. 1 in greater detail. The auxiliary switch is represented by a MOSFET 220 with its parasitic capacitances represented as gate to source Ciss1 242, source to drain Coss1 228 and gate to drain Crss1 240, and an inherent body diode 222 that is formed by the junction of an N Well with a P substrate or vice versa. The primary switch is represented by a MOSFET 224 with its parasitic capacitances represented as Ciss2 246, Coss2 248 and Crss2 244, and its inherent body diode 226. The power converter 200 further comprises a transformer 202, driving circuitry for the auxiliary switch 220 comprising three diodes 230, 232, 234 and a driving capacitor 236, and secondary rectification means comprising of a rectifier diode 210 and a smoothing capacitor 212.
The power converter 200 of FIG. 2 comprises a quasi resonant tank circuit 201. The first resonance capacitor 102 in FIG. 1 comprises the combination of the parasitic capacitor Coss1 228 and Crss1 240 of the auxiliary switch 220. Also, the parasitic capacitance Crss1 240 also appears in series with the second resonance capacitor 238 along with the capacitor Coss2 248 and capacitor Crss2 244 of the primary switch 242. The second resonance capacitor 106, of FIG. 1, is represented in FIG. 2 by the capacitor C2 238.
In most single ended power converters such as the flyback converters of FIGS. 1 and 2, it is desirable to keep a reset voltage limited so that the voltage level (Vds1) across a switching MOSFET such as the primary switch 224, in the case of the power converter remains within the safe operating area. In this situation, the reset voltage (Vres) across the primary winding of the transformer 202 is lower than the voltage across the primary winding of the transformer (Vde) during the on-time of the primary switch 224. To achieve Zero Voltage Switching for the primary switch 224, the energy (Ehigh) in the effective resonance capacitance at the point of maximum reset voltage (Vres) has to be equal or larger than the energy (Elow) in the effective resonance capacitance just prior to the switch on of the primary MOSFET. In general, as the operating frequency of a resonant capacitor approaches its self resonant frequency, the capacitive value will appear to increase causing an effective resonance capacitance that is higher than the physical capacitor's stated value.
A person of ordinary skill in the art will appreciate that the driving capacitor 236 is in parallel with the secondary resonant capacitor 238 when the auxiliary switch 220 is enabled. As a result, the values of the driving capacitor 236 and the secondary resonant capacitor 238 must be proportional; i.e. one should not be changed without changing the other since doing so would change the effective total resonance capacitance for the tank circuit 201. However, a capacitor of a smaller value cannot drive a capacitor of a higher value. As a result, in high power applications, such as power greater than 10 watts at an output, the auxiliary switch 220 must be larger in order to properly handle increased current flow. As the switch 220 increases in size, parasitic capacitances Crss1 and Ciss1 increase proportionally. As a result, the driving capacitor 236 must be of a higher value. A discrete capacitor of a higher value directly translates to increased size and cost, which can be unacceptable in small form factor products. Regardless of the size and cost, increasing all values may simply be untenable due to increased serial equivalent resistance of the capacitors causing unacceptable decreases in efficiency. To that end, what is needed is a power converter circuit having a quasi resonant tank circuit wherein the circuit driving the auxiliary switch is independent of the auxiliary switch itself.